Devices, base stations, and methods for communicating scheduling requests via an underlay control channel in a wireless communication system

ABSTRACT

Wireless communication systems, base stations, and user equipment are disclosed that enable communication of scheduling requests via an underlay control channel that has an energy below a noise level of the spectrum. The scheduling requests may be sent and received at any time, including during downlink and uplink data communication periods of the base station.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit under 35 U.S.C. §119(e) of U.S.Provisional Patent Application Ser. No. 62/455,364, filed Feb. 6, 2017,entitled “SYSTEMS, DEVICES, AND METHODS FOR COMMUNICATING SCHEDULINGREQUESTS VIA AN UNDERLAY CONTROL CHANNEL IN A WIRELESS COMMUNICATIONSYSTEM,” and U.S. Provisional Patent Application Ser. No. 62/372,611,filed Aug. 9, 2016, “SYSTEMS, DEVICES, AND METHODS FOR COMMUNICATINGSCHEDULING REQUESTS VIA AN UNDERLAY CONTROL CHANNEL IN A WIRELESSCOMMUNICATION SYSTEM,” the disclosure of each of which is herebyincorporated herein in its entirety by this reference.

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT

This invention was made with government support under Contract No.DE-AC07-05-ID14517 awarded by the United States Department of Energy.The government has certain rights in the invention.

FIELD

Embodiments of the present disclosure relate generally to systems,devices, and methods for communicating scheduling requests by userequipment via an underlay control channel, and receiving, processing,and granting the scheduling request by the base station.

BACKGROUND

There are two conventional approaches for user equipment (UE) initiateduplink (UL) transmission in a wireless system. In one approach the UEmay transmit a scheduling-request (SR) to the base station (BS), and inresponse the base station allocates UL resources and then informs the UEabout the allocated UL resources by transmitting a UL grant signal tothe UE. This type of UL resource allocation may be referred to as“scheduled-based UL access” (also referred to as the “scheduled-basedmethod” or a “grant-based method”). In a second approach, the UE maytransmit its data signals within an assigned set of UL resources that ispre-defined by the base station. This type of UL access may be referredto as “grant-free UL access” (also referred to as the “grant-freemethod”).

There is a trade-off between latency and resource utilization efficiencyin both the grant-free method and the scheduled-based method. In thegrant-free method, a pre-defined amount of resources is allocated with apre-defined periodicity. This type of allocation may have lower latencywith a cost of resource under-utilization if the UEs do not haveanything to transmit. On the other hand, the scheduled-based method mayimprove the resource-utilization efficiency with a cost of higherlatency as it takes multiple time-slots to receive the UL grants at theUE. As a consequence, both of these conventional methods may not besatisfactory for delay-sensitive applications (e.g., critical industrialcontrol, remote surgical robots, etc.) or other applications. Theseapplications are generically referred to as “ultra reliable low latencycommunication” (URLLC) in the industry.

BRIEF SUMMARY

In some embodiments, a user equipment device is configured to generate ascheduling request (SR) signal and transmit the SR signal to a basestation via an underlay control channel below a noise level for acommunication spectrum, and generate data packets and transmit the datapackets to the base station over an overlay channel above the noiselevel for the communication spectrum.

In some embodiments, a base station is configured to communicate uplinkdata and downlink data with a user equipment using active sub-carriersof a spectrum, and communicate scheduling requests with the userequipment using an underlay control channel below a noise level of thespectrum.

In some embodiments, a method comprises sending a scheduling requestfrom a user equipment to a base station via an underlay control channelbelow a noise level of a spectrum, receiving the scheduling request bythe base station, and transmitting a scheduling request grant from thebase station to the user equipment, wherein receiving the schedulingrequest and transmitting the scheduling request are performed inparallel with other data communication between the base station andother user equipment.

BRIEF DESCRIPTION OF THE DRAWINGS

The patent or application file contains at least one drawing executed incolor. Copies of this patent or patent application publication withcolor drawings will be provided by the Office upon request and paymentof the necessary fee. These drawings executed in color are found in theAppendices described below.

FIG. 1 is a schematic diagram of a wireless communication system.

FIG. 2 shows data streams for different base stations and one or moreUEs according to a conventional multi-cell synchronized TDD deployment.

FIG. 3 shows data streams between a base station and one or more UE,according to embodiments of the disclosure.

FIGS. 4 and 5 show data streams to illustrate UE initiated datatransmissions timeline comparisons between the conventional approach andembodiments of the disclosure.

FIGS. 6A, 6B, 6C show graphs that illustrate three options fortransmitting an SR signal while the base station is transmittingdownlink data to the UEs.

FIG. 7 is a wireless communication system illustrating an example forinterference caused by a nearby UE transmitting an SR signal whileanother UE is receiving the downlink data signal.

FIG. 8 is a wireless communication system illustrating a scenario forcalculating the probability of collision between an SR signal and adownlink data signal.

FIG. 9 illustrates subcarriers for SR signaling having a packet formataccording to an embodiment of the presented disclosure.

FIG. 10 is a graph presenting a packet error rate (PER) as a function ofthe receiver input signal-to-noise ratio (SNR) according to anembodiment of the present disclosure.

FIG. 11 is a simplified block diagram of the transmitter according to anembodiment of the disclosure.

FIG. 12 is a graph illustrating the PDFs of n_(i) and 1+n_(i) and thethreshold value that minimizes both the probability of misdetection andfalse alarm.

FIG. 13 are plots showing the probability distributions of thenormalized detector output {circumflex over (b)}_(i) for six choices ofSNR_(in).

FIG. 14 is a plot showing BER curves that compare the theoretical andsimulation results.

FIGS. 15 through 18 are plots showing curves for the probability ofmisdetection for the detectors according to various embodiments of thedisclosure.

DETAILED DESCRIPTION

In the following description, reference is made to the accompanyingdrawings in which are shown, by way of illustration, specificembodiments in which the disclosure may be practiced. The embodimentsare intended to describe aspects of the disclosure in sufficient detailto enable those skilled in the art to make, use, and otherwise practicethe disclosure. Furthermore, specific implementations shown anddescribed are only examples and should not be construed as the only wayto implement the present disclosure unless specified otherwise herein.It will be readily apparent to one of ordinary skill in the art that thevarious embodiments of the present disclosure may be practiced bynumerous other partitioning solutions. Other embodiments may be utilizedand changes may be made to the disclosed embodiments without departingfrom the scope of the disclosure. The following detailed description isnot to be taken in a limiting sense, and the scope of the presentinvention is defined only by the appended claims.

In the following description, elements, circuits, and functions may beshown in block diagram form in order not to obscure the presentdisclosure in unnecessary detail. Conversely, specific implementationsshown and described are exemplary only and should not be construed asthe only way to implement the present disclosure unless specifiedotherwise herein. Additionally, block definitions and partitioning oflogic between various blocks is exemplary of a specific implementation.It will be readily apparent to one of ordinary skill in the art that thepresent disclosure may be practiced by numerous other partitioningsolutions. For the most part, details concerning timing considerationsand the like have been omitted where such details are not necessary toobtain a complete understanding of the present disclosure and are withinthe abilities of persons of ordinary skill in the relevant art.

Those of ordinary skill in the art would understand that information andsignals may be represented using any of a variety of differenttechnologies and techniques. For example, data, instructions, commands,information, signals, bits, symbols, and chips that may be referencedthroughout the above description may be represented by voltages,currents, electromagnetic waves, magnetic fields or particles, opticalfields or particles, or any combination thereof. Some drawings mayillustrate signals as a single signal for clarity of presentation anddescription. It will be understood by a person of ordinary skill in theart that the signal may represent a bus of signals, wherein the bus mayhave a variety of bit widths, and the present disclosure may beimplemented on any number of data signals including a single datasignal.

The various illustrative logical blocks, modules, and circuits describedin connection with the embodiments disclosed herein may be implementedor performed with a general purpose processor, a special purposeprocessor, a Digital Signal Processor (DSP), an Application SpecificIntegrated Circuit (ASIC), a Field Programmable Gate Array (FPGA) orother programmable logic device, discrete gate or transistor logic,discrete hardware components, or any combination thereof designed toperform the functions described herein. A general-purpose processor maybe a microprocessor, but in the alternative, the processor may be anyconventional processor, controller, microcontroller, or state machine. Ageneral-purpose processor may be considered a special-purpose processorwhile the general-purpose processor executes instructions (e.g.,software code) stored on a computer-readable medium. A processor mayalso be implemented as a combination of computing devices, e.g., acombination of a DSP and a microprocessor, a plurality ofmicroprocessors, one or more microprocessors in conjunction with a DSPcore, or any other such configuration.

Also, it is noted that embodiments may be described in terms of aprocess that may be depicted as a flowchart, a flow diagram, a structurediagram, or a block diagram. Although a flowchart may describeoperational acts as a sequential process, many of these acts can beperformed in another sequence, in parallel, or substantiallyconcurrently. In addition, the order of the acts may be re-arranged. Aprocess may correspond to a method, a function, a procedure, asubroutine, a subprogram, etc. Furthermore, the methods disclosed hereinmay be implemented in hardware, software, or both. If implemented insoftware, the functions may be stored or transmitted as one or moreinstructions or code on computer-readable media. Computer-readable mediainclude both computer storage media and communication media, includingany medium that facilitates transfer of a computer program from oneplace to another.

It should be understood that any reference to an element herein using adesignation such as “first,” “second,” and so forth, does not limit thequantity or order of those elements, unless such limitation isexplicitly stated. Rather, these designations may be used herein as aconvenient method of distinguishing between two or more elements orinstances of an element. Thus, a reference to first and second elementsdoes not mean that only two elements may be employed there or that thefirst element must precede the second element in some manner. Inaddition, unless stated otherwise, a set of elements may comprise one ormore elements.

Embodiments of the disclosure include systems, devices, and methods inwhich UEs use an underlay channel to send a scheduling request to a basestation, which may result in reduced latency over conventional methods.Some embodiments may be included within a time division duplex (TDD)network deployment scenario. Other embodiments may be included within afrequency division duplex (FDD) deployments scenario. Embodiments arealso contemplated to be incorporated with various wireless communicationstandards including 4G LTE, 5G, and other future standards, as well asmachine-to-machine type communications, Internet of Things (IoT)applications, and other related applications.

Headings are included herein to aid in locating certain sections ofdetailed description. These headings should not be considered to limitthe scope of the concepts described under any specific heading.Furthermore, concepts described in any specific heading are generallyapplicable in other sections throughout the entire specification.

Deployment Scenario

FIG. 1 is a schematic diagram of a wireless communication system 100according to an embodiment of the disclosure. The wireless communicationsystem 100 includes a first base station 110A having a first coveragearea 112A that provides wireless communication services for UEs 120A₁,120A₂. The wireless communication system 100 also includes a second basestation 110B having a second coverage area 112B that provides wirelesscommunication services for UEs 120B₁, 120B₂. In some embodiments, UEs120A₁, 120A₂, UEs 120B₁, 120B₂ may operate in half-duplex mode, and thebase stations 110A, 110B may operate in the full-duplex (FD) mode withself-interference cancellation capabilities. Each of the base stations110A, 120B and the UEs 120A₁, 120A₂, 120B₁, 120B₂ may include componentssuch as one or more processors, transmitters, receivers, memory, etc.configured to enable each device to perform the functions describedherein.

FIG. 2 shows data streams 200 for different base stations (e.g., BS1,BS2) and one or more UEs according to a conventional multi-cellsynchronized TDD deployment having time slots k, k+1, k+2, k+3, and soon. At the beginning of each time-slot, each base station transmits itsdownlink control information (DCI) 202 to the UEs within its coveragearea. Each base station may then transmit either downlink (DL) data 204or receive uplink (UL) data 206 from the associated UEs, after which thebase station receives UL control information (UCI) 208 from the UEs.Each of the DCI 202, DL data 204, UL data 206, and UCI 208 are multipleusers access channels.

FIG. 3 shows data streams 300 between a base station and one or more UEaccording to embodiments of the disclosure. In particular, theembodiment of FIG. 3 shows a dynamic TDD deployment in which the UEsinterested in transmitting UL data to the base station transmit ascheduling-request (SR) signal 306A, 306B, 306C to the base stationduring the DL data 302 or UL data 304 time-slots. The SR signals 306A,306B, 306C may be non-orthogonal multiple access underlay controlchannel (UCC) based signals. Throughout the disclosure, this UCC-basedSR signal may be referred to as PSRUCH (Physical LayerScheduling-Request Underlay Channel). In some embodiments, the UCC maybe always on coexisting with the overlay signals (e.g., active band)available for communicating SR signals. As a result, there does not needto be any assignment of resources by the base station in order to make ascheduling request. Receiving and granting scheduling requests may bedone in parallel with other communications rather than in sequence,which may reduce latency in setting up data communication between the UEand the base station.

In some embodiments, the UCC may use the entire UCC bandwidth by using aspread spectrum signaling method described in U.S. Pat. No. 8,731,027,issued May 20, 2014, U.S. Pat. No. 8,861,571, issued Oct. 14, 2014, andU.S. Pat. No. 9,369,866, issued Jun. 14, 2016, the entire disclosure ofeach which is incorporated herein by this reference. Other underlaysignaling methods are also contemplated in embodiments of thedisclosure.

Referring again to FIG. 3, the UEs may transmit the SR signals 306A,306B, 306C using PSRUCH communications such that they are received atthe base station below the noise level. The SR signals 306A, 306B, 306Cmay contain control information, such as buffer status, priority,power-headroom, etc., related to the data packets to be transmitted bythe UEs. The serving base station decodes the SR signals 306A, 306B,306C and uses the information to schedule the UEs in the upcomingtime-slots. Based on the number of SR signals 306A, 306B, 306C beingtransmitted, the base station may have the flexibility to adapt bydeferring some of the delay-tolerant scheduled UL transmissions andallocate UL resources and transmit UL grants in response to the receivedSR signals 306A, 306B, 306C.

A dedicated common control channel is available for the SR signals;however, as mentioned earlier, a dynamic resource allocation scheme mayprovide a capability to access resources “on-demand” in order to meetthe latency requirements. PSRUCH may offer that capability in a moreefficient manner by not requiring any dedicated resources and providingan “always-on” availability for the contention-based SR transmissions,allowing the UEs to access UL resources with very low latency andsignaling overhead.

While embodiments described herein primarily refer to TDD deployments,it should be recognized that FDD deployments are also contemplated inembodiments of the disclosure. In some FDD deployments, PSRUCH may beapplied only in the uplink carrier to provide the benefits as in the TDDdeployment scenario. FD operation mode may transmit and receivesimultaneously on a common carrier, in comparison with half duplex modewhere transmission and reception occur in different time-slots on acommon carrier. In some embodiments, Transmit Receive Points (TRPs) withthe base station controller in the network cloud may also be used fordeployment scenarios.

FIGS. 4 and 5 show data streams 400, 500 to illustrate UE initiated datatransmissions timeline comparisons between the conventional approach(FIG. 4) and embodiments of the disclosure (FIG. 5). According to Table1 below, the SR-related latency in the UL is the dominant component ofthe total delay using conventional methods.

TABLE 1 Time Component Description (ms) 1 Average waiting time forPUCCH - Physical 5/0.5 Uplink Control Channel - (10 ms/1 ms SR period) 2UE sends Scheduling Request on PUCCH 1 3 Base station decodes SR andgenerates 3 the Scheduling Grant 4 Transmission of the Scheduling Grant1 5 UE processing Delay (decoding of scheduling 3 grant + L1 encoding ofUL data) 6 Transmission of UL data 1 7 Data decoding in base station 3Total delay (ms) 17/12.5

As shown in FIG. 4, in order to send an SR the UE must wait for anSR-valid UL resource 408 at the end of a time slot. This wait time afterinitial data arrival during DL data 404, on average, could vary from 0.5ms to 5 ms assuming 1 ms and 10 ms SR resource periods, respectively.The base station may then decode and generate the scheduling grant anentire time slot later when it sends its DCI 402 transmission before ULdata 406 may be sent.

On the other hand, as shown in FIG. 5, sending SR signals 510 overPSRUCH enables the UEs to transmit SR signals 510 immediately after dataarrival such that in response the base station can transmit thecorresponding UL grants in the very next DCI 502 transmission. Thisscheme may reduce the average waiting time to almost zero for the UE totransmit its SR signal 510 after data arrival and also reduces the delayuncertainty.

The UE initiated data transmissions using PSRUCH may be desirable fordelay-sensitive type data transmissions requiring low latency andflexibility in data formats. On the other hand, conventionalscheduling-based schemes and the grant-free UE transmissions may beoperable for the delay-tolerant scheduled and small-packet size typedata transmissions, respectively.

Interference Analysis PSRUCH Transmissions During the Transmission ofthe Control Information Channels

The delivery of the DCI and UCI should be reliable and robust.Therefore, it is desirable that interference avoidance is achievedduring the DCI and UCI transmissions. As a consequence, the PSRUCHtransmissions may not be allowed during the transmission of the DCI.However, during UCI, PSRUCH transmission may be allowed. This is becausewith a reasonable power control, PSRUCH may be guaranteed to remainbelow the noise floor of the UCI signals received at the base station.

PSRUCH Transmissions During the Downlink Data Transmissions

Scheduling cannot be used without some interference to the downlinktransmissions because SR signals are random transmissions using PSRUCH.FIGS. 6A, 6B, 6C show graphs 600A, 600B, 600C that illustrate threeoptions for transmitting an SR signal while the base station istransmitting downlink data to the UEs.

FIG. 6A shows the UCC signals 606A including the SR signals being spreadover the entire bandwidth of the active subcarriers 602 and the guardband 604 as part of the UCC below the noise level. FIG. 6B shows the UCCsignals 606B including the SR signals being spread over a more limitedportion of the bandwidth of the spectrum (e.g., confined to be withinthe guard bands 604) but still as part of the UCC below the noise level.By being limited to the guard bands 604, the power used to transmit theUCC signals 606B may be greater, if desired, than the power of the UCCsignals 606A of FIG. 6A that are spread across the entire bandwidth.FIG. 6C show transmission of signals 606C as a narrowband signal withinthe guard bands 604. This situation may not be desirable forasynchronous contention-based uplink signals, such as thescheduling-request signals or in some cases where guard bands 604 maynot be available (e.g., guard bands used for LTE NB-IoT Rel-13 basedservices, etc.).

FIG. 7 is a wireless communication system 700 illustrating an examplefor interference I_(P) caused by a nearby UE (e.g., UE 120A₂)transmitting an SR signal while another UE (e.g., UE 120A₁, 120B₁) isreceiving the downlink data signal. Interference I_(s) is theself-interference. In order to reduce interference, power control andvarious spreading techniques may be considered to keep the SR signalsbelow the noise level at the receivers (e.g., by using a long PN-codesequence spread along both time and frequency). With such well-spreadsignals the interference caused by the far-off UEs transmitting SRsignals via PSRUCH may be substantially mitigated due to thepropagation-loss. However, some of the UEs may not be able to decodetheir respective DL signal in the presence of a nearby UE transmittingan SR request. In this case, the base station 110 may schedule a hybridautomatic repeat request (HARQ) retransmission of the DL data signal tothe specific UEs that were unable to decode the prior received DL datatransmissions. HARQ retransmissions may be reduced to an acceptablelevel by proper spreading of the SR signal over a sufficiently largenumber of subcarriers and across time.

Power control may be applied to the PSRUCH transmissions of the SRsignals such that the SR signals reach the base station 110 receiver atbelow the noise level. In addition, if needed, the self-interferenceI_(s) may be suppressed using various interference cancellation schemesknown in the art. Both the power control and the self-interferencesuppression may contribute to detecting and decoding the received PSRUCHsignals at the base station 110.

PSRUCH Transmissions During the Uplink Data Transmissions

At both the serving and the non-serving base station receivers thePSRUCH signals are received below the noise-level causing no significantinterference. After demodulating the uplink data signals, the servingbase station receiver simply de-spread the PSRUCH signals to obtain thescheduling-request information. Note PSRUCH has an advantage compared tothe conventional uplink control SR signaling such as the LTE PUCCHbecause PSRUCH requires no spectrum fragmentation, no dedicatedresources and it is always available to the UEs.

Probability of Collision of SR Signal with Downlink Data Signal at aNearby UE

To show the collisions of PSRUCH with downlink signals has a lowprobability, here, we analyze a simplified network to confirm that suchcollisions have low probability of occurrence.

FIG. 8 is a wireless communication system 800 illustrating a scenariofor calculating the probability of collision between PSRUCH and adownlink data signal. A single cell of the wireless communication system800 is shown in which the base station 110A (also referred to as “BS”)at the center of the cell 112A. The first user equipment 120A₁ (alsoreferred to as “UE₁”) is transmitting a PSRUCH signal. At the same time,a second UE 120A₂ (also referred to as “UE₂”) is receiving a DL signalfrom the base station 110A. When the PSRUCH signal has a destructiveinterference (I) with the DL signal at the second UE 120A₂, a collisionhas occurred.

The channel additive noise is assumed to originate mostly from thethermal noise. For the base station 110A, which is operating in a fullduplex mode, it is assumed a well-designed interference cancellation isdeployed. p may be used to denote the power spectral density (PSD) ofthe channel noise. It is assumed that power control is used to determinethe PSD of all the transmit signals. For normal UL and DL signals, thePSD of the received signals is assumed to be α dB above the noise level.For the PSRUCH signal, the PSD of the received signals is assumed to beβ dB below the noise level. The PSRUCH signal will collide with the DLsignal at the second UE 120A₂, if it introduces interference at a levelγ dB or greater less than the DL received signal. To simplify thecalculations here, only the line-of-sight (LOS) path loss is consideredfor all the communications and use the following equations:

P _(loss)=(89.5+16.9 log₁₀ d) dB   (1)

where d is the distance between the transmitter and receiver.

To proceed, r may be used to denote the distance between the basestation 110A and the first UE 120A₁ and d_(i) to denote the distancebetween the UEs 120A₁, 120A₂. Following the statements made above fordifferent signal levels:

PSD of PSRUCH signal at BS=ρ−β dBm   (2)

Using Equation (2) and considering Equation (1):

$\begin{matrix}{{{PSD}\mspace{14mu} {of}\mspace{14mu} {PSRUCH}\mspace{14mu} {signal}\mspace{14mu} {at}\mspace{14mu} {UE}_{2}} = {\rho - \beta - {16.9\; \log_{10}\frac{d_{i}}{r - d_{i}}{dBm}}}} & (3)\end{matrix}$

and:

PSD of DL signal at UE₂=ρ+α dBm.   (4)

A collision may occur at UE₂ when:

PSD of DL signal at UE₂−PSD of PSRUCH signal at UE₂≦γ.   (5)

Using Equations (3) and (4) in (5), and assuming that d_(i)<<r:

$\begin{matrix}{{\alpha + \beta + {16.9\; \log_{10}\frac{d_{i}}{r}}} \leq \gamma} & (6)\end{matrix}$

Rearranging Equation (6) results in:

d_(i)≦ηr   (7)

where:

$\begin{matrix}{\eta = 10^{\frac{\gamma - \alpha - \beta}{16.9}}} & (8)\end{matrix}$

The above results tell us a collision occurs when UE₂ is within a diskof radius ηr from UE₁. Assuming that UE₁ and UE₂ are uniformlydistributed over a cell disk with the radius of R, the probability ofUE₁ being at a distance r from the center of the cell is obtained as:

$\begin{matrix}{{p(r)} = {\frac{2\pi \; r}{\int_{0}^{R}{2\pi \; {rdr}}} = \frac{2\; r}{R^{2}}}} & (9)\end{matrix}$

Using this result, the average area of the collision disk (the disk withthe radius ηr) is obtained as:

Average area of the collision disk=∫₀ ^(R) π(ηr)2p(r)dr=πη ² R ²   (10)

Dividing this average area by the total area of the cell, πR², leads to:

Collision Probability=η²   (11)

To give a sense of how small or large is this probability of collision,consider the case where α=20 dB, β=10 dB, and γ=10 dB. For thisscenario, we get collision probability=0.0043. This is a very smallprobability and, thus, quite acceptable in a network with HARQcapability.

Embodiments of the present disclosure may also use PSRUCH for pagingapplications. Paging is an occasional and periodic transmission that isoften used by the base station to inform UEs within its coverage area ofchanges in the network. During the paging periods (also referred to as“paging occasions”), the UEs that are in IDLE state (e.g., sleep mode)temporarily wake up and listen to the paging channel. If the UE detectsits own identity in the received information then the UE processes thecorresponding downlink paging message and the scheduling information toset itself up to receive the assigned data channel. Otherwise, if the UEis not paged, the UE may return to its IDLE state until the next pagingoccasion or other usage.

Using PSRUCH, the base station may page its UEs using an UCC. In such anembodiment, the underlay paging channel may be transmitted using eitherof underlay transmission options FIG. 6A or FIG. 6B while the UEstransmit the uplink data signals. One advantage of this paging scheme isthe base station may not be required to allocate predefined resourcesfor the paging channel. The underlay transmission of the page via theunderlay communication channel may also require full-duplex capable UEsbecause some UEs may be transmitting an uplink signal while receivingthe page from the base station.

Additional examples for implementation of the UCC SR channel arepresented and evaluated below. It should be understood, however, thatthe following implementations described herein are non-limitingexamples, and that additional variations of UCC SR channels arecontemplated and within the scope of the disclosure.

FIRST EXAMPLE OF UCC SR CHANNEL: AN OFDM SIGNALING METHOD

Embodiments of the disclosure include a packet format for UCC SRchannel. In some embodiments, OFDM signaling may be employed as in LTEwith similar subcarrier spacing. As an example, a subcarrier spacing of15 kHz and a bandwidth of 18 MHz available for UCC transmission mayaccommodate 1200 subcarriers. Each SR packet may include two OFDMsymbols. The first OFDM symbol may be used for channel estimation andcarry pilot symbols. The first OFDM symbol may be shared among a numberof users (e.g., K users). In such a case, a subset of subcarriers may beallocated to each user for channel estimation. For example, if K=3 usersshare the same OFDM symbol for channel estimation, subcarrier numbers 1,4, 7, . . . may be allocated to user one, subcarrier numbers 2, 5, 8, .. . may be allocated to user 2, and subcarrier numbers 3, 6, 9, . . .may be allocated to user 3. The second OFDM symbol of each user may beused to carry the information bits of that user.

FIG. 9 illustrates subcarriers for SR signaling having a packet formataccording to an embodiment of the present disclosure. In this example,three users U₁, U₂, U₃ share one OFDM symbol for channel estimation, andeach user U₁, U₂, U₃ may be assigned one OFDM symbol for informationtransmission. As shown in FIG. 9, four OFDM symbols S₁, S₂, S₃, and S₄are used to serve users U₁, U₂, U₃. OFDM symbol S₁ may be used totransmit pilot symbols from each of the three users. OFDM symbol S₂ maybe used to transmit information from the first user U₁. OFDM symbol S₃may be used to transmit information from the second user U₂. OFDM symbolS₄ may be used to transmit information symbols from the third user U₃.

In some embodiments, each OFDM information symbol may carry a total of12 uncoded bits. These 12 bits may be expanded to 36 coded bits andmapped to 18 QPSK symbols. Each QPSK symbol may be spread across

$\frac{1200}{18} \approx 16$

subcarriers that are spaced out across the full bandwidth of 18 MHz. Forcoding, a rate ⅓ convolutional code may be used, similar to transmissionof control information (PUCCH/PUSCH) in LTE.

UEs may be carrier and time synchronized with the base station. As aresult, the cyclic prefix striped samples of the OFDM symbols S₁, S₂,S₃, and S₄ are available to the receiver (e.g., base station) forprocessing. Channel estimation may be performed using the maximumlikelihood estimator (MLE) as described in M. Morelli and U. Mengali, “Acomparison of pilot-aided channel estimation methods for OFDM systems,”in IEEE Transactions on Signal Processing, vol. 49, no. 12, pp.3065-3073, Dec 2001, the disclosure of which is hereby incorporatedherein in its entirety by this reference. The estimated channel may beused for despreading of the demodulated data symbols through a set ofmaximum ratio combiners. The results of this stage may be passed to aViterbi decoder for information recovery.

FIG. 10 is a set of graphs 1000 presenting the frame error rates (FERs)as a function of the receiver input signal-to-noise ratio (SNR)according to an embodiment of the present disclosure. The channel noisemay be Gaussian white noise. The FERs are presented in FIG. 10 fordifferent choices of the parameter K (number of users). As seen, whenone user (i.e., K=1) fully uses one OFDM symbol for channel estimationand uses one OFDM symbol for information transmission, a FER of lowerthan 10⁻² at SNR of −15 dB may be achieved as shown by curve 1012. Thismeans more than 99% of SR packets may be successfully decoded when theSR packets are transmitted at a level −15 dB below the signal activitiesin the channel.

FIG. 10 shows additional cases where multi-users share the same OFDM forchannel estimation and different OFDM symbols for data transmission asdiscussed above and visualized in FIG. 9. The FERs for the case of K=2,3, 4, and 5 users are curves 1014, 1018, 1024, 1032. As seen, for thesecases, a performance loss of about 1 dB may be achieved when K isincremented by one unit. Other cases presented in FIG. 10 include thesituation in which multi-users share the same OFDM symbol for channelestimation and more than one OFDM symbols are shared for datatransmissions. For instance, the case (3, 2) shown by curve 1020 meansthree users share one OFDM symbol for channel estimation and two OFDMsymbols are shared for transmissions of the data of the three users. Insuch cases a multiuser technique is used for joint detection of theinformation of the users.

SECOND EXAMPLE OF UCC SR CHANNEL: ON/OFF KEYING USING ZC SEQUENCES

Embodiments of the present disclosure may include an on/off keyingtechnique configured to send a single-bit of information from atransmitter (e.g., mobile terminal) to receiver (e.g., a base station).By using each bit with a given signature (i.e., a spreading gain vector)to transmit information to the base station, a variety of messages maybe transmitted by multiple UEs by employing a sufficient number ofspreading gain vectors. Each information bit may include a distinctspreading gain vector.

The number of single-bit messages that may be transmitted varies withthe available resources. Some limitations may be imposed by the channeland the target signal to interference plus noise ratio (SINR) that areneeded for reliable transmission. Embodiments of the disclosure mayinclude an on/off keying method that includes Zadoff-Chu (ZC) sequencesin time domain as spreading gain vectors is a direct sequence spreadspectrum system. The ZC sequences, as further explained below, may allowconstruction of different message signals with guaranteed nointer-message interference, even in the presence of a channel.Transmission of multiple messages from the same UE may be permitted.Therefore, the SR messages sent from one user may imply a differentrequest, for example, to send SR messages for different packet sizes. Ifthree packet sizes are used, each UE may be assigned three different ZCsequences. In that case, the transmission of each of the three ZCsequences may imply SR messages for one of the three packet sizes.

Certain properties of the ZC sequences may be used to develop asignaling method for the desired on/off keying method. A relativelylarge number of independent on/off keying bits (e.g., in the order of 20to 50 bits) may be transmitted within each OFDM symbol while theorthogonality of the respective bits may be within a satisfactoryapproximation. That is, the bits may be transmitted without substantialinterference with each other. Given that in each subframe there may be 7OFDM symbols, this method allows transmission of 140 to 350 independentbits in each subframe. If each UE is assigned 3 bits, this method maysupport up to

$\frac{350}{3} \approx {120\mspace{14mu} {{UEs}.}}$

In case shorter subframes are adopted (e.g., mini-subframes eachcarrying 2 OFDM symbols), some embodiments may be limited to support asmaller number of UEs. Conversely, a larger number of UEs may besupported for embodiments in which longer subframes are adopted. In someembodiments, a much larger number of UEs may be served through asignaling method in which each bit is transmitted twice. An example ofsuch an embodiment is discussed below in the section “ON/OFF KeyingContinued: Receiver Implementation and Theoretical Results (CSIunknown).” In particular, three variations of the receiverimplementation are discussed below under the subsections “FirstDetector,” “Second Detector,” and “Third Detector,” as non-limitingexamples of receivers that may be implemented according to embodimentsof the disclosure.

As an example, a ZC sequence of length N may be represented by thecolumn vector:

$\begin{matrix}{z_{0} = \begin{bmatrix}z_{0} \\z_{1} \\z_{2} \\\vdots \\z_{N - 1}\end{bmatrix}} & (12)\end{matrix}$

The circularly shifted versions of z₀ may be defined as z₁, z₂, . . . ,z_(N-1):

$\begin{matrix}{{z_{1} = \begin{bmatrix}z_{N - 1} \\z_{0} \\z_{1} \\\vdots \\z_{N - 2}\end{bmatrix}},{z_{2} = \begin{bmatrix}z_{N - 2} \\z_{N - 1} \\z_{0} \\\vdots \\z_{N - 3}\end{bmatrix}},\ldots \mspace{14mu},{z_{N - 1} = \begin{bmatrix}z_{1} \\z_{2} \\z_{3} \\\vdots \\z_{0}\end{bmatrix}}} & (13)\end{matrix}$

The on/off keying method may build upon the following property of the ZCsequences. The vectors z₀, z₁, z₂, . . . , z_(N−1) are orthogonal toeach other. As a result, for all values of i and j:

z_(i) ^(H)z_(j)=0, for any i ≠ j,   (14)

where the superscript H denotes Hermitian (i.e., conjugate transpose).

To transmit an SR bit, a ZC sequence vector z_(i) may be treated as thespreading gain vector of a CDMA transmitter (i.e., form a directsequence spread spectrum (DS-SS) signal), and accordingly transmit asingle bit b=+1. Moreover, to allow orthogonality of different bits, theCDMA method implemented may be CP-CDMA. In other words, cyclic prefix(CP) is added to each CDMA symbol.

FIG. 11 is a simplified block diagram of the transmitter 1100 accordingto an embodiment of the disclosure. The transmitter 1100 may beconfigured to transmit a DS-SS signal pre-pended with a cyclic prefix.As shown in FIG. 11, the CP-assisted DS-SS signal is generated such thatit will be aligned with the OFDM symbols within the respective cell. Inother words, the DS-SS signals that are adopted are compatible with OFDMsignals within the network, thus, can benefit from any synchronizationthat has been adopted. The spreading sequence is a cyclic-shiftedZC-sequence. Different numbers of shifts may be applied for each bit toassure orthogonality of the signals associated with different UEs ordifferent bits that are associated with the same UE. A subset of Mcyclic-shifted sequences out of total N cyclic-shifted versions of theroot ZC-sequence are assigned to each UE if there are M different sizesof uplink transmission formats available. For example, in the URLLCcase, three different uplink transmission format sizes (e.g., 32, 50,200 bytes) may be considered. Accordingly, three cyclic-shiftedZC-sequences may be assigned to a UE.

In addition, to allow simple synchronization of the transmitted SR bitswithin a network, each CDMA bit may have the same length as the OFDMsymbols in the network, and the generated CP-CDMA bits are time-alignedwith the OFDM symbols. To develop the background theory that enables theorthogonality of the SR bits, without any loss of generality, the ZCsequence z_(i) is transmitted as an SR message. After passing throughthe channel, and removing the cyclic prefix at the receiver, thereceived signal vector may be:

x_(i)=Z_(i)h_(i)   (15)

where h_(i) is a vector of length L representing the channel impulseresponse and Z_(i) is a matrix of size N×L and with the columns z_(i),z_(i+1), . . . , z_(i+L−1). If another bit is sent simultaneously withthe spreading gain vector z_(j), the respective received signal may be:

x_(j)=Z_(j)h_(j)   (16)

Recalling the orthogonality property of Equation (14), when |i−j|>L, thereceived signal vectors x_(i) and x_(j) are orthogonal to each other,i.e., x_(i) ^(H)x_(j)=0.

Within each OFDM symbol, it is possible to transmit up to N/L bits, froma single UE or from different UEs, without interfering with one another.With seven OFDM symbols within each downlink subframe, a total of

$\frac{7\; N}{L}$

bits may be transmitted over each subframe. We also note that the valueof L that enables inter-bit interference free transmission may be equalto the length of the cyclic prefix. In other words, the cyclic prefixlength may be set equal to L. With a typical cyclic prefix length of 7%of the OFDM symbol length N,

$\frac{7\; N}{L} = {\frac{7\; N}{0.07\; N} = 100}$

UEs can transmit their underlay SR signals per subframe-duration withoutinterfering with each other. As a result, within each OFDM subframe 100inter-bit interference free bits may be transmitted.

Furthermore, typical channel responses have usually relatively long lowenergy tails. As a result, L (i.e., the spacing between ZC sequences)may be half of the cyclic prefix length or smaller at a cost ofintroducing a small amount of inter-bit interference. Moreover, becauseonly a small subset of UEs transmit SR message in a given subframe, thechance of presence of such interference is relatively low and, thus, onemay argue for further reduction of the parameter L. Reducing L willallow SR services be expanded to a larger number of UEs, or a largernumber of bits may be transmitted by each user. Although the parameter Lmay be made smaller than the cyclic prefix to serve a larger number ofUEs in some embodiments, for purposes of discussion, it is assumedherein that L is equal to CP length to simplify the discussion.

ON/OFF Keying Continued: Receiver Implementation and Theoretical Results(CSI Known)

Analysis and numerical results that are presented in this section maynot achievable in practice as channel state information (CSI) is unknownin practice and its estimation may require some resources and are notperfect. Hence, the disclosure serves to demonstrate a potential upperlimit to the performance of at least some of the embodiments of thedisclosure.

As an example, in the scenario with two UEs in the network, the receivedsignal x within one OFDM time frame may be:

x=b _(i) x _(i) +b _(j) x _(j) +n   (17)

where b_(i) and b_(j) are the transmitted bits and n represents othersignal activities (including channel noise) within the transmissionband.

Assuming that x_(i) and x_(j) are known (i.e., Z_(i) and Z_(j) and therespective channel impulse responses h_(i) and h_(j) are known), and theelements of n are a set of independent and identically distributed(i.i.d.) random variables, the detectors for estimation of b_(i) andb_(j) may a pair of matched filters expressed as:

d_(i)=x_(i) ^(H)x   (18)

and

d_(j)=x_(j) ^(H)x.   (19)

Such detectors may not be practical for implementation, as h_(i) andh_(j) may be unknown to the receiver. Also, in practice, the elements ofn may not be independent. However, understanding of the performance ofdetectors like Equation (18) and Equation (19) still may be of interest,as they provide some intuition to what one should expect from a gooddetector. For this detector, next, its processing gain is evaluated.

Processing Gain of the Detector

Consider the detector represented by Equation (18). SubstitutingEquation (17) in Equation (18) and recalling that x_(i) ^(H)x_(j)=0, thedetector may be represented as:

d _(i) =b _(i) x _(i) ^(H) x _(i) +x _(i) ^(H) n   (20)

Because d_(i) is independent of the signal part that carries b_(j) andrecalling (18), for the UE that has transmitted b_(i), we define the SNRat the receiver input as:

$\begin{matrix}{{SNR}_{i} = {\frac{x_{i}^{H}x_{i}}{E\left\lbrack {n^{H}n} \right\rbrack} = \frac{h_{i}^{H}Z_{i}^{H}Z_{i}h_{i}}{N\; \sigma_{n}^{2}}}} & (21)\end{matrix}$

where σ_(n) ² is the variance of the elements of n. Assuming that the ZCvectors are normalized to the length of unity, Equation (21) may besimplified as:

$\begin{matrix}{{SNR}_{i} = {\frac{h_{i}^{H}h_{i}}{N\; \sigma_{n}^{2}}.}} & (22)\end{matrix}$

At the detector output, Equation (20) implies that the magnitude of thedetected bit has the known energy x_(i) ^(H)x_(i)=h_(i) ^(H)h_(i). Theproduct h_(i) ^(H)h_(i) may be a constant independent of the channelh_(i), which implies that the bit power at the detector output (assumingthat a bit has been transmitted) is equal to (h_(i) ^(H)h_(i))². On theother hand, the noise power at the detector output may be obtained byevaluating

$\frac{1}{2}{{E\left\lbrack {x_{i}^{H}{nn}^{H}x_{i}} \right\rbrack}.}$

In this example, a factor of ½ has been added because the detected bitsare real-valued and, hence, only the real part of the noise at thedetector output may be accounted.

When the elements of n are samples of a white noise,

${\frac{1}{2}{E\left\lbrack {x_{i}^{H}{nn}^{H}x_{i}} \right\rbrack}} = {\frac{1}{2}\sigma_{n}^{2}h_{i}^{H}h_{i}}$

and, thus, the SNR at the detector output is obtained as:

$\begin{matrix}{{SNR}_{o} = {\frac{\left( {h_{i}^{H}h_{i}} \right)^{2}}{\frac{1}{2}\sigma_{n}^{2}h_{i}^{H}h_{i}} = {\frac{2\left( {h_{i}^{H}h_{i}} \right)}{\sigma_{n}^{2}} = {2N \times {{SNR}_{i}.}}}}} & (23)\end{matrix}$

Equation (23) shows a processing gain of 2N. When the elements of n aresamples of a bandlimited process, Equation (23) may be modified. Inspread spectrum systems, in general, the processing gain varies with thebandwidth. Thus, when the elements of n are samples of a process that isspread over a fraction of the full band, the processing gain 2N reducesto 2βN. As a result, Equation (23) may be modified as:

SNR_(o)=2βN×SNR_(i),   (24)

wherein the processing gain of the proposed underlay channeldecreases/increases with the bandwidth of transmission.

Probability of Misdetection and False Alarm

With the detector setup discussed above, the output of the detectorafter normalization with the constant term E[h_(i) ^(H)h_(i)] and takingthe real part yields the following form:

{circumflex over (b)} _(i) =b _(i) +n _(i)   (25)

where n_(i) is a real-valued Gaussian term with variance:

$\begin{matrix}{\sigma_{n_{i}}^{2} = {\frac{1}{{SNR}_{o}} = \frac{1}{2\beta \; N \times {SNR}_{i}}}} & (26)\end{matrix}$

When b_(i)=0 (i.e., no SR has been sent) {circumflex over(b)}_(i)=n_(i), and for a correct detection that b_(i)=0, the real partof n_(i) may be below a set threshold. On the other hand, when b_(i)=1(i.e., an SR has been sent) {circumflex over (b)}_(i)=1+n_(i), and for acorrect detection that b_(i)=1, the real part of 1+n_(i) may be abovethe same threshold.

FIG. 12 is a graph 1200 illustrating the probability distributionfunctions (PDFs) of n_(i) (line 1210) and 1+n_(i) (line 1220) and thethreshold value that minimizes both the probability of misdetection andfalse alarm. As shown in FIG. 12, setting a threshold (line 1215) at ½minimizes both the probability of misdetection (i.e., detecting no SRwhen an SR has been sent) and the probability of false alarm (i.e.,detecting an SR when no SR has been sent). Moreover, with the setthreshold these two probabilities are equal. These two probabilities maybe combined as one and called the bit-error-rate (BER).

With the above observations, the bit-error-rate (BER) may be calculatedusing the following:

$\begin{matrix}{{BER} = {Q\left( \frac{1}{2\sigma_{n_{i}}} \right)}} & (27)\end{matrix}$

where

${Q(x)} = {\frac{1}{\sqrt{2\pi}}{\int_{x}^{\infty}{e^{{- u^{2}}/2}{{du}.}}}}$

To evaluate Equation (27) the Q-function may be replaced by thecomplementary error function, which results in:

$\begin{matrix}{{BER} = {\frac{1}{2}{{erfc}\left( \frac{1}{2\sqrt{2}\sigma_{n_{i}}} \right)}}} & (28)\end{matrix}$

Computer Simulations

FIG. 13 are plots 1300 showing the probability distributions of thenormalized detector output {circumflex over (b)}_(i) for six choices ofSNR_(in). The theoretical prediction curves that use Equation (28) forcomputation of the BER (lines 1302/1306, 1312/1316, 1322/1326,1332/1336, 1342/1346, 1352/1356) in each case are compared withrespective computer simulations (lines 1304/1308, 1314/1318, 1324/1328,1334/1338, 1344/1348, 1354/1358) that show a substantial match betweeneach comparison.

FIG. 14 is a plot 1400 showing BER curves 1402-1408 that compare thetheoretical and simulation results. The results in FIG. 14 are presentedfor two choices of ZC sequence length (i) N₁=1024 and bandwidth 15 MHz;and (ii) N₂=512 and bandwidth 7.5 MHz. In some embodiments, zeros may beinserted between ZC-sequences samples to be expanded and match the FFTlength of N=1024 and N=2048 for 10 MHz and 20 MHz channel bandwidth,respectively. Coincidentally, the plot 1400 also indicates theprobability of false alarm. The number of simultaneous UEs that can besupported per symbol-duration depends upon the length of the ZC-sequenceafter expansion, N, and the length of CP.

Balancing Between the Probability of Correct Detection and False AlarmRate

The threshold value that is shown in FIG. 12 may equalize theprobability of a false alarm (i.e., detecting an ON bit when no bit hasbeen transmitted) and the probability of a misdetection (i.e., notdetecting an ON bit). To achieve a balance between these twoprobabilities may include moving the threshold line 1215 in FIG. 12 tothe left or to the right. By moving the threshold line 1215 to the leftthe probability of false alarm increases while the probability ofmisdetection decreases. On the other hand, if the threshold line 1215 isshifted to the right, the probability of false alarm decreases while theprobability of misdetection increases. Assuming, the variance of n_(i)is known, setting the threshold line 1215 for a fixed false alarm rate,the probability of correct detection may improve as SNR increases.

ON/OFF Keying Continued: Receiver Implementation and Theoretical Results(CSI Unknown)

For the received signal given by Equation (17), it was assumed x_(i) wasknown. This requires the knowledge of h_(i), which is usuallyunavailable. Thus, it may be desirable to estimate the value of b_(i) inthe absence of a knowledge of h_(i). The knowledge regarding h_(i) maybe limited to its length of L and possibly the fact that only a few ofthe elements of h_(i) are significantly different from zero (i.e., h_(i)is sparse). Under this condition, Equation (17) may be rewritten as:

x=b _(i) Z _(i) h _(i) +b _(j) x _(j) +n   (29)

In Equation (29), x and Z_(i) may be known and that Z_(i) ^(H)x_(j)=0.Based on this knowledge, it may be desirable to determine the results ifb_(i)=0 or 1. To this end, a length L vector may be constructed as:

y_(i)=Z_(i) ^(H)x

y _(i) =b _(i) h _(i) +n′ _(i)   (30)

where n′_(i)=Z_(i) ^(H)n.

At least some knowledge of the noise statistics may be required for areasonable detector design that is able to detect of the value of b_(i)from the vector y_(i). In Equation (30), the vector n′_(i) is a Gaussianvector (when n is a Gaussian noise) or, in general, may be approximatedby a Gaussian vector. To make the development and analysis of thedesired detector convenient, n′_(i) may be assumed to be a Gaussianvector. Also, assuming that the elements of the vector n areuncorrelated with one another, the element of the vector n′_(i) may alsobe uncorrelated. Hence, because n′_(i) is assumed to be Gaussian, itsstatistics may be known if its variance could be estimated.

To allow estimation of the variance of the elements of n′_(i) anddetection of the presence or absence of an information bit, informationbit b may be assumed to be transmitted twice, through twonon-overlapping ZC sequences. Accordingly, instead of Equation (30), thefollowing pair of equations are assumed to be available at the receiver:

$\begin{matrix}\left\{ \begin{matrix}{y_{1} = {{bh} + n_{1}^{\prime}}} \\{y_{2} = {{bh} + n_{2}^{\prime}}}\end{matrix} \right. & (31)\end{matrix}$

In Equation (31) the subscript “i” is removed from all the variables forbrevity of the derivations that follow.

Noise Variance Estimation

In Equation (31), the noise vectors n′₁ and n′₂ are assumed to have thesame statistics, but are independent of each other. With thisassumption, to obtain an estimate of the variance σ_(n) ² of theelements of these noise vectors, it is noted that:

Δy=y ₁ −y ₂ =n′ ₁ −n′ ₂   (32)

and, accordingly, obtain:

$\begin{matrix}{\sigma_{n}^{2} = \frac{\left( {\Delta \; y} \right)^{H}\left( {\Delta \; y} \right)}{2L}} & (33)\end{matrix}$

where L is the length of Δy (or, equivalently, the length of h) and thefactor of 2 in the denominator of Equation (33) comes from the fact thateach element of Δy is a summation of two noise terms.

Embodiments of the disclosure may include detectors that take y₁, y₂,and σ_(n) ² as inputs and generate a decision value {circumflex over(b)} at its output. Three contemplated detectors are discussed below;however, additional detectors are also contemplated. The false alarmrate may be held constant for these detectors while letting theprobability of misdetection to improve (e.g., decrease) as SNRincreases.

First Detector

For the first detector, construction of the decision variable mayinclude:

d ₁ =

{y ₁ ^(H) y ₂ }=bh ^(H) h+

{n′ ₁ ^(H) n′ ₂ +bh ^(H) n′ ₂ +n′ ₁ ^(H) h}  (34)

where

{x} means the real part of x. Also, in some of the subsequent equations,ℑ{x} is used to denote the imaginary part of x.

When b=0, Equation (34) reduces to:

x=d ₁|_(b=0) =

{n′ ₁ ^(H) n′ ₂}.   (35)

When σ_(n) ² is known, a threshold η may be selected that satisfies:

P(x>η)=p_(FA)   (36)

for a predetermined probability of false alarm, p_(FA).

Derivation of an equation that enables Equation (36) to be solved for ηwhen p_(FA) is specified leads to the following probability densityfunction (PDF) for the random variable x that is defined in (35).

$\begin{matrix}{{f_{1}(x)} = \frac{\left( \frac{2{x}}{\sigma_{n}^{2}} \right)^{L - \frac{1}{2}}{K_{L - \frac{1}{2}}\left( \frac{2{x}}{\sigma_{n}^{2}} \right)}}{\frac{\sigma_{n}^{2}}{2}2^{L - \frac{1}{2}}\pi^{\frac{1}{2}}{\Gamma (L)}}} & (37)\end{matrix}$

where Γ(·) is the Gamma function and K_(a)(·) is the a-order modifiedBessel function of second kind. For an integer argument L,Γ(L)=L!=L(L−1)(L−2) . . . 1. The a-order modified Bessel function ofsecond kind is expressed as:

$\begin{matrix}{{K_{a}(x)} = {\frac{\pi}{2} \cdot \frac{{I_{- a}(x)} - {I_{a}(x)}}{\sin \left( {a\; \pi} \right)}}} & (38)\end{matrix}$

where I_(a)(x) is the modified Bessel function of first kind, expressedas:

$\begin{matrix}{{I_{a}(x)} = {\sum\limits_{m = 0}^{\infty}{\frac{1}{{m!}{\Gamma \left( {m + a + 1} \right)}}\left( \frac{x}{2} \right)^{{2m} + a}}}} & (39)\end{matrix}$

Using Equation (37), for a given value of p_(FA), the threshold η isfound by solving the following equation, for the unknown η:

1−F ₁(η)=p _(FA)   (40)

where:

F ₁(η)=∫_(−∞) ^(η) f ₁(x)dx   (41)

is the cumulative distribution function (CDF) of the random variable x.

Here, the solution to Equation (40) is found numerically, by firstmaking a plot of p_(FA)=1−F₁(η) as a function of η, from which for agiven p_(FA), the threshold η can be found.

Second Detector

For the second detector, the averaged signal vector may be defined as:

$\begin{matrix}{y_{3} = \frac{y_{1} + y_{2}}{2}} & (42)\end{matrix}$

This averaged vector y₃ may have the same form as y₁ and y₂, with 3 dBSNR improvement. The total energy of the averaged vector y₃ (i.e., thedecision variable) may be evaluated as:

d₂=y₃ ^(H)y₃   (43)

and based on the result one may make a decision about the value of b. Tothis end, Equation (41) implies:

y ₃ =bh+n′ ₃   (44)

where n′₃ is a complex-valued Gaussian random vector with variance

$\frac{\sigma_{n}^{2}}{2}$

at each of its elements. When b=0, Equation (43) reduces to:

x=d ₂|_(b=0) =

{n′ ₃ ^(H) n′ ₃}.   (45)

Here, x is a χ² random variable with 2 L degrees of freedom and, thus,with the PDF of:

$\begin{matrix}{{f_{2}(x)} = {\frac{1}{{\Gamma (L)}\sigma_{n}^{2L}}x^{L - 1}e^{- \frac{x}{\sigma_{n}^{2}}}}} & (46)\end{matrix}$

Next, starting with Equation (46), the threshold η can be found bymaking a plot of p_(FA)=1−F₂(η) as a function of η, from which for agiven p_(FA), the threshold can be found. Here:

F ₂(η)=∫_(−∞) ^(η) f ₂(x)dx   (47)

Third Detector

For the third detector, the impulse response h may be sparse and themagnitude of the real and imaginary parts of the impulse response h aresignificantly larger than σ_(n) in the range of interest where theprobability of misdetection is low. With this property of the impulseresponse h in mind, the following decision variables may be defined:

Type 1:

$\begin{matrix}{{d_{3,1} = {\max {z_{1}}}},{z_{1} = \begin{bmatrix}{\Re \left\{ y_{1} \right\}} \\{\left\{ y_{1} \right\}} \\{\Re \left\{ y_{2} \right\}} \\{\left\{ y_{2} \right\}}\end{bmatrix}}} & (48)\end{matrix}$

Type 2:

$\begin{matrix}{{d_{3,2} = {\max {z_{2}}}},{z_{2} = \begin{bmatrix}{\Re \left\{ y_{3} \right\}} \\{\left\{ y_{3} \right\}}\end{bmatrix}}} & (49)\end{matrix}$

In Equation (48) and Equation (49), the absolute value operation |·| isapplied to z₁ and z₂ elementwise.

For Type 1, when b=0, z₁ is a vector of length 4 L with independentGaussian entries, each with variance of

$\frac{\sigma_{n}^{2}}{2}.$

Accordingly, tor a given threshold

$\begin{matrix}{p_{FA} = {{P\left( {{{d_{3,1} > \eta}b} = 0} \right)} = {1 - \left( {{erf}\left( \frac{\eta}{\sigma_{n}} \right)} \right)^{4L}}}} & (50)\end{matrix}$

In Equation (50), the error function

${erf}\left( \frac{\eta}{\sigma_{n}} \right)$

is the probability of each of the elements of z₁ remain smaller than η.If all 4 L elements of z₁ are smaller than η, there will be no falsealarm, otherwise a false alarm happens, i.e., b=1 is detected, while bhas been equal to zero.

For Type 2, when b=0, z₂ is a vector of length 2 L with independentGaussian entries, each with variance of

$\frac{\sigma_{n}^{2}}{4}.$

Accordingly, for a given threshold η:

$\begin{matrix}{p_{FA} = {{P\left( {{{d_{3,2} > \eta}b} = 0} \right)} = {1 - \left( {{erf}\left( \frac{\eta \sqrt{2}}{\sigma_{n}} \right)} \right)^{2L}}}} & (51)\end{matrix}$

Computer Simulations

The above detectors for three different choices of the channel and twochoice of p_(FA)=0.01 and 0.001 may be evaluated with respect to theresults shown in FIGS. 15 through 18. The channels used are the TDLmodels A, B, and C that have been suggested by the 3GPP LTE group forfuture communications with carrier frequencies above 6 GHz. Inparticular, FIG. 15 is a plot 1500 showing curves 1502-1512 for theprobability of misdetection for the first detector, FIG. 16 is a plot1600 showing curves 1602-1612 for the probability of misdetection forthe second detector, FIG. 17 is a plot 1700 showing curves 1702-1712 forthe probability of misdetection for the third detector (Type 1), andFIG. 18 is a plot 1800 showing curves 1802-1812 for the probability ofmisdetection for the third detector (Type 2).

In FIGS. 15 through 18 the probability of misdetection refers to thecase that a bit b=1 has been transmitted and the receiver has failed todetect the bit. The results are presented for two choices of p_(FA)=0.01and 0.001. As seen the variation of the results with the change inchannel and p_(FA) is relatively small, particularly, for the first andthe second detectors. For these detectors we see a variation of about 1dB. For the third detector, this variation increases to about 2 dB.Moreover, the first detector and second detector show a superiorperformance over the third detector.

Beside the probability of misdetection, another consideration inselecting the detector may be the number of independent bits/UEs that adetector can serve over each ZC (or each OFDM) symbol duration. Asdiscussed above, for a ZC sequence of length Nand a channel duration Lsamples, M=N/L orthogonal bits may be transmitted over each ZC symbolduration. With repetition of each bit twice, it may appear that thenumber of orthogonal bits will be halved. While this may be the case,with a choice of a proper detector, the introduced interference may betolerated if one of the repetitions of a pair of bits overlap. If suchinterference ignored, the number of orthogonal bits that can betransmitted over each ZC symbol duration may be

$\begin{pmatrix}M \\2\end{pmatrix} = {\frac{M!}{{\left( {M - 2} \right)!}{2!}} = {\frac{M\left( {M - 1} \right)}{2}.}}$

In certain LTE scenarios, M=N/L may be as large as 50, hence, the numberof independent bits that can be transmitted may be as large as

$\frac{5049}{2} = 1225$

in such an example. As a non-limiting example, if 6 variations of SRsare allocated to each UE, such a set up can serve as many as 200 UEs ina cell. Among the three detectors discussed above, the first detectormay introduce a minimum inter-bit interference.ON/OFF Keying with Multiple Bits

The ON/OFF keying method discussed above transmits a signal (an SR)whose presence or absence is detected at the receiver (e.g., the basestation of a cell). In some embodiments, the following modification canbe applied to the generated SR signal such that multiple bits aretransmitted per SR. An ON SR will consist of a reference bit b₀=1 andmultiple bits that are transmitted over a set of non-overlapping ZCsequences. As a non-limiting example, two information bits b₁ and b₂taking the values of ±1 may be transmitted. Under this condition,equation (31) extends to:

$\begin{matrix}\left\{ \begin{matrix}{y_{0} = {h + n_{0}^{\prime}}} \\{y_{1} = {{b_{1}h} + n_{1}^{\prime}}} \\{y_{2} = {{b_{2}h} + n_{2}^{\prime}}}\end{matrix} \right. & (52)\end{matrix}$

The first line of equation (52) is a reference signal and the next twolines are the information bit carriers. Following the first detector ofthe previous section, here, the presence of the SR is assumed if atleast two of the following inequalities are held:

$\begin{matrix}\left\{ \begin{matrix}{{{\Re \left\{ {y_{0}^{H}y_{1}} \right\}}} > \eta} \\{{{\Re \left\{ {y_{0}^{H}y_{2}} \right\}}} > \eta} \\{{{\Re \left\{ {y_{1}^{H}y_{2}} \right\}}} > \eta}\end{matrix} \right. & (53)\end{matrix}$

Other methods are contemplated to similarly detect the presence of theSR. Once the presence of the SR is detected, the estimates of theinformation bits are obtained as:

$\begin{matrix}\left\{ \begin{matrix}{{\hat{b}}_{1} = {{sgn}\left\lbrack {\Re \left\{ {y_{0}^{H}y_{1}} \right\}} \right\rbrack}} \\{{\hat{b}}_{2} = {{sgn}\left\lbrack {\Re \left\{ {y_{0}^{H}y_{2}} \right\}} \right\rbrack}}\end{matrix} \right. & (54)\end{matrix}$

where “sgn” refers to the signum of the indicated variable.

Embodiments of the disclosure may be extended to transmission of morethan two bits of information should be apparent to one of ordinary skillin the art.

While the present disclosure has been described herein with respect tocertain illustrated embodiments, those of ordinary skill in the art willrecognize and appreciate that it is not so limited. Rather, manyadditions, deletions, and modifications to the illustrated embodimentsmay be made without departing from the scope of the disclosure ashereinafter claimed, including legal equivalents thereof. In addition,features from one embodiment may be combined with features of anotherembodiment while still being encompassed within the scope of thedisclosure. Further, embodiments of the disclosure have utility withdifferent and various detector types and configurations.

What is claimed is:
 1. A user equipment device configured to: generate ascheduling request (SR) signal and transmit the SR signal to a basestation via an underlay control channel below a noise level for acommunication spectrum; and generate data packets and transmit the datapackets to the base station over an overlay channel above the noiselevel for the communication spectrum.
 2. The user equipment device ofclaim 1, wherein the user equipment device is configured to not allowcommunication over the underlay control channel while receiving downlinkcontrol information from the base station.
 3. The user equipment deviceof claim 1, wherein the user equipment device is configured to allowcommunication over the underlay control channel while transmittinguplink control information to the base station.
 4. The user equipmentdevice of claim 1, wherein the user equipment device is configured totransmit the SR signal via an active underlay control channel while theuser equipment is also communicating with the base station via an activeband overlay signal.
 5. The user equipment device of claim 1, whereinthe underlay control channel is a multiple access underlay controlchannel.
 6. The user equipment device of claim 1, wherein the SR signalincludes control information related to the data packets to betransmitted by the UEs, wherein the control information includes atleast one of a buffer status, priority information, power-headroom, andcombinations thereof.
 7. A base station configured to communicate uplinkdata and downlink data with a set of user equipment using activesub-carriers of a spectrum, and communicate scheduling requests with theuser equipment using an underlay control channel below a noise level ofthe spectrum.
 8. The base station of claim 7, wherein the schedulingrequests are spread spectrum signals spread across the entire spectrumbelow the noise level.
 9. The base station of claim 7, wherein thescheduling requests are spread spectrum signals spread below the noiselevel and across a portion the of spectrum within guard bands outside ofthe active subcarriers.
 10. The base station of claim 7, wherein thebase station is configured to receive the scheduling requests from theUE without first assigning uplink or downlink resources to the UE. 11.The base station of claim 7, wherein the base station is configured tocommunicate the uplink data, the downlink data, and the schedulingrequests with the user equipment within a time division duplex (TDD)network or a frequency division duplex (FDD) network.
 12. The basestation of claim 7, wherein the based station is further configured togenerate and transmit paging messages to the user equipment using theunderlay control channel.
 13. A method comprising: sending a schedulingrequest from a user equipment to a base station via an underlay controlchannel below a noise level of a spectrum; receiving the schedulingrequest by the base station; and transmitting a scheduling request grantfrom the base station to the user equipment, wherein receiving thescheduling request and transmitting the scheduling request are performedin parallel with other data communication between the base station andother user equipment.
 14. The method of claim 13, wherein sending theother data communication between the base station and the other userequipment includes downlink data communication and/or uplink datacommunication between the base station and other user equipment.
 15. Themethod of claim 14, wherein the downlink data communication and/or theuplink data communication is transmitted over an overlay channel abovethe noise level for the communication spectrum.
 16. The method of claim13, further comprising: receiving a plurality of different schedulingrequests from different user equipment via the underlay control channelby the base station at different times; and transmitting schedulingrequest grants for each of the different scheduling requests to theirrespective user equipment.
 17. The method of claim 13, wherein sendingthe scheduling requests includes using an on/off keying techniqueconfigured to send information from the UE to the base station.
 18. Themethod of claim 17, wherein each information bit sent from the UE to thebase station includes a distinct spreading gain vector.
 19. The methodof claim 18, wherein the spreading gain vector includes a cyclic-shiftedZC-sequence as its spreading sequence.
 20. The method of claim 17,wherein each scheduling request includes a reference bit and multiplebits over a set of non-overlapping ZC sequences per transmittedscheduling request.